OPA686 Burr-Brown, OPA686 Datasheet - Page 13

• Manufacturer Part Number: OPA686
• Description: Wideband / Low Noise / Voltage Feedback OPERATIONAL AMPLIFIER
• Manufacturer: Burr-Brown

output power and frequency. Finally, the distortion increases
as the fundamental frequency increases due to the rolloff in
the loop gain with frequency. Conversely, the distortion will
improve going to lower frequencies down to the dominant
open-loop pole at approximately 100kHz. Starting from the
–82dBc 2nd harmonic for a 5MHz, 2Vp-p fundamental into
a 200
Curves), the 2nd harmonic distortion for frequencies lower
than 100kHz will be approximately –82dBc – 20log(5MHz/
100kHz) = –116dBc.
The OPA686 has extremely low 3rd-order harmonic distor-
tion. This also gives a high two-tone, 3rd-order
intermodulation intercept as shown in the Typical Perfor-
mance Curves. This intercept curve is defined at the 50
load when driven through a 50
direct comparisons to RF MMIC devices. This matching
network attenuates the voltage swing from the output pin to
the load by 6dB. If the OPA686 drives directly into the input
of a high impedance device, such as an ADC, the 6dB
attenuation is not taken. Under these conditions, the inter-
cept will increase by a minimum 6dBm. The intercept is
used to predict the intermodulation spurious for two, closely-
spaced frequencies. If the two test frequencies, f
specified in terms of average and delta frequency, f
(f
spurious tones will appear at f
between two equal test-tone power levels and these
intermodulation spurious power levels is given by
the Typical Performance Curve and P
dBm at the 50
frequencies. For instance, at 5MHz the OPA686 at a gain of
+10 has an intercept of 48dBm at a matched 50 load. If the
full envelope of the two frequencies needs to be 2Vp-p, this
requires each tone to be 4dBm. The 3rd-order intermodulation
spurious tones will then be 2 • (48 – 4) = 88dBc below the
test-tone power level (–84dBm). If this same 2Vp-p, two-
tone envelope were delivered directly into the input of an
ADC—without the matching loss or the loading of the 50
network—the intercept would increase to at least 54dBm.
With the same signal and gain conditions, but now driving
directly into a light load, the spurious tones will then be at
least 2 • (54 – 4) = 100dBc below the 4dBm test-tone power
levels centered on 5MHz.
DC ACCURACY AND OFFSET CONTROL
The OPA686 can provide excellent DC signal accuracy due
to its high open-loop gain, high common-mode rejection,
high power supply rejection, and low input offset voltage
and bias current offset errors. To take full advantage of its
low 1.5mV input offset voltage, careful attention to input
bias current cancellation is also required. The low noise
input stage of the OPA686 has a relatively high input bias
current (10 A typical into the pins) but with a very close
match between the two input currents—typically 100nA
input offset current. The total output offset voltage may be
reduced considerably by matching the source impedances
looking out of the two inputs. For example, one way to add
dBc = 2 • (IM3 – P
1
+ f
2
)/2 and f = |f
load at G = +10 (from the Typical Performance
load for one of the two closely-spaced test
O
) where IM3 is the intercept taken from
2
– f
1
|/2, the two 3rd-order, close-in
O
matching resistor to allow
3 •
O
is the power level in
f. The difference
1
and f
2
, are
O
=
13
bias current cancellation to the circuit of Figure 1 would be
to insert a 20
from the 50
resistor is DC-coupled, this will increase the source resis-
tances for the non-inverting input bias current to 45 . Since
this is now equal to the resistance looking out of the
inverting input (R
the bias currents to the output leaving only the offset current
times the feedback resistor as a residual DC error term at the
output. Using the 453
will now be less than 0.9 A • 453 = 0.4mV over the full
temperature range.
A fine-scale, output offset null, or DC operating point
adjustment, is often required. Numerous techniques are
available for introducing a DC offset control into an op amp
circuit. Most of these techniques eventually reduce to setting
up a DC current through the feedback resistor. One key
consideration to selecting a technique is to insure that it has
a minimal impact on the desired signal path frequency
response. If the signal path is intended to be non-inverting,
the offset control is best applied as an inverting summing
signal to avoid interaction with the signal source. If the
signal path is intended to be inverting, applying the offset
control to the non-inverting input can be considered. For a
DC-coupled inverting input signal, this DC offset signal will
set up a DC current back into the source that must be
considered. An offset adjustment placed on the inverting op
amp input can also change the noise gain and frequency
response flatness. Figure 8 shows one example of an offset
adjustment for a DC-coupled signal path that will have
minimum impact on the signal frequency response. In this
case, the input is brought into an inverting gain resistor with
the DC adjustment an additional current summed into the
inverting node. The resistor values setting this offset adjust-
ment are much larger than the signal path resistors. This will
insure that this adjustment has minimal impact on the loop
gain and hence, the frequency response.
FIGURE 8. DC-Coupled, Inverting Gain of –20, with
10k
+5V
–5V
5k
5k
Output Offset Adjustment.
terminating resistor. When the 50
series resistor into the non-inverting input
0.1µF
F
V
|| R
0.1µF
I
G
20k
), the circuit will cancel the gains for
feedback resistor, this output error
50
R
48
G
OPA686
±200mV Output Adjustment
OPA686
V
V
+5V
–5V
O
I
= –
1k
Supply Decoupling
R
F
R
R
G
Not Shown
F
= –20
source
V
O
®