OPA631N BURR-BROWN [Burr-Brown Corporation], OPA631N Datasheet - Page 13

• Manufacturer Part Number: OPA631N
• Description: Low Power, Single-Supply OPERATIONAL AMPLIFIERS TM
• Manufacturer: BURR-BROWN [Burr-Brown Corporation]

BANDWIDTH VS GAIN: NON-INVERTING OPERATION
Voltage feedback op amps exhibit decreasing closed-loop
bandwidth as the signal gain is increased. In theory, this
relationship is described by the Gain Bandwidth Product
(GBP) shown in the specifications. Ideally, dividing GBP by
the non-inverting signal gain (also called the Noise Gain, or
NG) will predict the closed-loop bandwidth. In practice, this
only holds true when the phase margin approaches 90 , as it
does in high gain configurations. At low gains (increased
feedback factors), most amplifiers will exhibit a more com-
plex response with lower phase margin. The OPA631 and
OPA632 are compensated to give a slightly peaked response
in a non-inverting gain of 2 (Figure 1). This results in a
typical gain of +2 bandwidth of 75MHz, far exceeding that
predicted by dividing the 68MHz GBP by 2. Increasing the
gain will cause the phase margin to approach 90 and the
bandwidth to more closely approach the predicted value of
(GBP/NG). At a gain of +10, the 7.6MHz bandwidth shown
in the Typical Specifications is close to that predicted using
the simple formula and the typical GBP.
The OPA631 and OPA632 exhibit minimal bandwidth re-
duction going to +3V single supply operation as compared
with +5V supply. This is because the internal bias control
circuitry retains nearly constant quiescent current as the total
supply voltage between the supply pins is changed.
INVERTING AMPLIFIER OPERATION
Since the OPA631 and OPA632 are general purpose,
wideband voltage feedback op amps, all of the familiar op
amp application circuits are available to the designer. Figure
5 shows a typical inverting configuration where the I/O
impedances and signal gain from Figure 1 are retained in an
inverting circuit configuration. Inverting operation is one of
the more common requirements and offers several perfor-
mance benefits. The inverting configuration shows improved
slew rate and distortion. It also allows the input to be biased
at V
be independently moved to be within the output voltage
range with coupling capacitors, or bias adjustment resistors.
In the inverting configuration, three key design consider-
ation must be noted. The first is that the gain resistor (R
becomes part of the signal channel input impedance. If input
impedance matching is desired (which is beneficial when-
ever the signal is coupled through a cable, twisted pair, long
PC board trace or other transmission line conductor), R
may be set equal to the required termination value and R
adjusted to give the desired gain. This is the simplest
approach and results in optimum bandwidth and noise per-
formance. However, at low inverting gains, the resultant
feedback resistor value can present a significant load to the
amplifier output. For an inverting gain of 2, setting R
50
requires a 100
advantage that the noise gain becomes equal to 2 for a 50
source impedance—the same as the non-inverting circuits
considered above. However, the amplifier output will now
see the 100
S
/2 without any headroom issues. The output voltage can
for input matching eliminates the need for R
feedback resistor in parallel with the external
feedback resistor. This has the interesting
M
G
but
G
to
G
F
)
13
FIGURE 5. Gain of –2 Example Circuit.
load. In general, the feedback resistor should be limited to
the 200
increase both the R
then achieve the input matching impedance with a third
resistor (R
the parallel combination of R
The second major consideration, touched on in the previous
paragraph, is that the signal source impedance becomes
part of the noise gain equation and hence influences the
bandwidth. For the example in Figure 5, the R
combines in parallel with the external 50
ance, yielding an effective driving impedance of 50
576
for calculating the noise gain. The resultant is 2.87 for
Figure 5, as opposed to only 2 if R
discussed above. The bandwidth will therefore be lower for
the gain of –2 circuit of Figure 5 (NG = +3) than for the
gain of +2 circuit of Figure 1.
The third important consideration in inverting amplifier
design is setting the bias current cancellation resistors on
the non-inverting input (a parallel combination of R
263 ). If this resistor is set equal to the total DC resistance
looking out of the inverting node, the output DC error, due
to the input bias currents, will be reduced to (Input Offset
Current) • R
in Figure 5, the total resistance to ground between the
inverting input and the source will be 401 . Combining
this in parallel with the feedback resistor gives the R
263
frequency noise introduced by this resistor, and power
supply feedthrough, R
long as R
As a minimum, the OPA631 and OPA632 require an R
Source
50
= 26.8 . This impedance is added in series with R
used in this example. To reduce the additional high
0.1 F
T
M
to 1.5k
< 400 , its noise contribution will be minimal.
) to ground. The total input impedance becomes
F
57.6
. If the 50
R
374
M
R
G
523
523
2R
2R
F
OPA631, OPA632
and R
T
T
range. In this case, it is preferable to
T
is bypassed with a capacitor. As
G
source impedance is DC-coupled
values as shown in Figure 5, and
OPA63x
+5V
G
and R
750
M
R
F
could be eliminated as
M
0.1 F
.
DIS
source imped-
50
+
R
O
6.8 F
50
M
Load
value
T
T
G
=
=
||
T
®